1. Field of the Invention
The present invention relates to an amplifier as well as a transmitter and a communication device incorporating the same. More particularly, the present invention relates to an amplifier which operates with a high frequency, as well as a transmitter and a communication device incorporating the same.
2. Description of the Background Art
A power amplifier to be used in a transmission block of a mobile phone or the like must be able to amplify a signal on the order of several W with a high gain.
FIG. 10 is a diagram illustrating an exemplary structure of a conventional amplifier 900 designed for amplifying a weak received signal (see, for example, Behzad Razavi, “RF Microelectronics”, Prentice-Hall, Nov. 1, 1997, p.169).
In the amplifier 900 shown in FIG. 10, a signal which is input via a terminal 901 is applied to a base of a transistor 910. The signal is amplified, led through a collector, and taken out at an output terminal 902. To the base, a bias Vb is applied via a resistor 921. To the collector, a bias Vc is applied via an inductor 922. In order to ensure separation between the collector bias and the base bias, a coupling capacitor 920 which exhibits a sufficiently small impedance at a fundamental frequency fc is connected at the base side.
Between the base and the collector of the transistor 910, there exists a parasitic capacitor 923 (having a capacitance Cp). An inductor 924 (having a reactance Lx) is connected to the parasitic capacitor 923 in parallel. It is assumed herein that the reactance Lx of the inductor 924 is chosen so that parallel resonance occurs with the parasitic capacitor 923 at the frequency fc at which amplification is expected, and that eq. 1 below is satisfied at the frequency fc (angular frequency ω=2πfc):Lx×Cp=1/ω2  eq. 1.
By prescribing such a reactance value for the inductor 924, the inductor 924 and the parasitic capacitor 923 function so that a parallel resonance occurs between the base and the collector, thus maximizing the impedance. Since power feedback is suppressed as a result of this, a decrease in the gain of the transistor is reduced. By thus reducing the decrease in the gain of the transistor, it becomes possible to reduce the decrease in the power added efficiency (PAE) of the amplifier.
FIG. 11 is a diagram illustrating an exemplary structure of a transmitter 930 which performs a polar coordinate modulation (Polar Modulation) operation to substantially improve the operation efficiency of an amplifier (see, for example, U.S. Pat. No. No. 6,256,482). In the transmitter 930 shown in FIG. 11, a digital baseband signal which has been subjected to a π/4 shift QPSK(Quadrature Phase Shift Keying) modulation is input to a signal generation section 931, which is composed of a DSP(Digital Signal Processor) and a DAC(Digital to Analog Converter). The signal generation section 931 extracts a phase component from the digital baseband signal, and outputs it as a phase control signal Eph(t). Moreover, the signal generation section 931 extracts an amplitude component from the digital baseband signal, and outputs it as an amplitude modulation signal Emag(t).
The phase control signal Eph(t) is input to a quadrature modulator 932. Based on the phase control signal Eph(t), the quadrature modulator 932 modulates a carrier signal having a frequency f0, and outputs a phase modulated signal on which the phase control signal Eph(t) is superposed, such that the phase control signal Eph(t) has a certain envelope. A filter 933 removes any unwanted signal component from the phase modulated signal. The output signal from the filter 933 is input to a three-staged power amplification block including a driver-stage amplifier 934, an inter-stage amplifier 935, and a last-stage amplifier 936, the three stages being in a serial connection in this order.
The driver-stage amplifier 934 has a certain voltage being supplied to its power terminal 937 from a battery. The driver-stage amplifier 934 amplifies the phase modulated signal, and inputs the amplified signal to the inter-stage amplifier 935.
On the other hand, the amplitude modulation signal Emag(t) is amplified by a high-efficiency Class D amplifier 938. The filter 939 removes any unwanted component from the signal output signal from the Class D amplifier 938, and outputs the resultant signal as an amplitude modulation signal Efm(t). The amplitude modulation signal Efm(t) is split into two portions. That is, the amplitude modulation signal Efm(t) is input to a power supply section (not shown) of the inter-stage amplifier 935 and to a power supply section (not shown) of the last-stage amplifier 936.
In accordance with the amplitude modulation signal Efm(t) which is input to the power supply section, the inter-stage amplifier 935 amplifies or attenuates the phase modulated signal which is output from the driver-stage amplifier 934.
The last-stage amplifier 936 mixes the phase modulated signal which is output from the inter-stage amplifier 935 and the amplitude modulation signal Efm(t) which is input to its power supply section, and thus outputs a π/4 shift QPSK modulated signal having been superposed on the carrier signal. It is assumed that the envelope of the π/4 shift QPSK modulated signal has a voltage Eo(t).
Thus, a phase modulated signal and an amplitude modulation signal are generated from a digital baseband signal, and the amplitude modulation signal having been input to the power supply section of the last-stage amplifier 936 is mixed with the phase modulated signal. In this manner, a signal which has been subjected to a polar coordinate modulation (Polar Modulation) in accordance with the digital baseband signal is obtained as an output.
Now, problems associated with the amplifier shown in FIG. 10 will be described. In practice, the transistor not only includes a parasitic capacitor, but also inherently includes a base-collector capacitance (referred to as an “intrinsic capacitor”) within the transistor.
FIG. 12 is a schematic diagram illustrating an equivalent circuit of the transistor 910 shown in FIG. 10. A resistor 912 and a capacitor 913 exist between the base and the emitter. A constant-current source 914 and an output resistance 915 exist between the collector and the emitter. An intrinsic capacitor (base-collector capacitance) 911 exists between the base and the collector.
In the case of an amplifier for transmission signals, unlike in an amplifier for received signals, a “large-sized” transistor is generally required in order to be able to output a high-power signal. As a result of this, the intrinsic capacitor 911 has a large capacitance, and the capacitance of the intrinsic capacitor 911 becomes more dominant than is the capacitance of the parasitic capacitor 923 as shown in FIG. 10. Therefore, merely providing the inductor 924 which resonates with the parasitic capacitor 923 shown in FIG. 10 (as is described in the aforementioned document authored by Behzad Razavi) causes a shift in the resonance frequency, thereby making it difficult to sufficiently prevent a gain decrease at the desired frequency (hereinafter this problem will be referred to as “the first problem”).
In the communication techniques of the recent years, it is a requirement for the amplifier to have a uniform gain over a wide frequency range. In the amplifier 900 shown in FIG. 10, the Q value of the parallel resonance circuit composed of the inductor 924 and the parasitic capacitor 923 is increased due to the low resistance component. This high Q value means that the impedance of the resonance circuit may greatly vary depending on the frequency, thus preventing the amplifier from having a uniform gain over a wide frequency range (hereinafter this problem will be referred to as “the second problem”).
Next, problems associated with the transmitter shown in FIG. 11 will be described. First, the operation of the last-stage amplifier 936 will be specifically described. In general, a transmitter in a mobile phone must be able to control the output power level depending on the distance from a base station. Hereinafter, such control of output power will simply be referred to as “power control”. In order to be attain such power control, the transmitter must have a wide dynamic range. Herein, a case will be discussed where the average output power of the output modulated signal needs to be varied by a range of 20 dBm, from 10 dBm to 30 dBm.
FIG. 13 is a graph illustrating the relationship between the source voltage Vcc and the output voltage Vo in the last-stage amplifier 936 shown in FIG. 11. Granted that the last-stage amplifier 936 is always operating in a saturation region, basically, the output voltage Vo increases in proportion with the source voltage Vcc, as shown in the area enclosed by dotted lines in FIG. 13.
The π/4 QPSK modulated signal is a signal whose amplitude varies with time. For a given average power, the amplitude of the π/4 QPSK modulated signal has a dynamic range of about 18 dB, from +3 dB to −15 dB.
By allowing the amplitude modulation signal Efm(t) to vary within range A of source voltage Vcc, a modulated signal having a maximum average power of 30 dBm can be output. Within range A, the output voltage Vo varies linearly with respect to the source voltage Vcc. Therefore, as accurate amplitude information which is in proportion with the amplitude modulation signal Efm(t), the last-stage amplifier 936 outputs a π/4 shift QPSK modulated signal having the envelope Eo(t).
On the other hand, by allowing the amplitude modulation signal Efm(t) to vary within range B of source voltage Vcc, a modulated signal having a minimum average power of 10 dBm can be output. Within range B, the output power Vo does not vary linearly with respect to the source voltage Vcc linearly. Therefore, the output modulated signal is distorted, i.e., the envelope Eo(t) has inaccurate amplitude information. This results in a deteriorated communication quality.
Next, the reason why the output voltage Vo does not vary linearly with respect to the source voltage Vcc within range B will be described.
In order for the output voltage Vo to vary linearly, the last-stage amplifier 936 must be operating entirely within the saturation region.
Regions X and Y in FIG. 13 are regions in which the output voltage Vo does not vary linearly because the last-stage amplifier 936 is not operating within the saturation region.
In the region X, the input power is small relative to the source voltage Vcc, and therefore the last-stage amplifier 936 does not saturate. Thus, the output voltage Vo does not vary despite the increasing source voltage Vcc.
The source voltage of the inter-stage amplifier 935 varies so as to result in the same potential as that defining the source voltage of the last-stage amplifier 936. Therefore, in the region Y, the inter-stage amplifier 935 cannot provide output power which is sufficient for allowing the last-stage amplifier 936 to saturate adequately.
A region Z is a region in which the output voltage Vo does not vary linearly although the last-stage amplifier 936 is operating within the saturation region. In the region Z, the input power is very large relative to the source voltage Vcc. Therefore, power may leak to the output side via the parasitic capacitor and the intrinsic capacitor in the last-stage amplifier 936, which explains the reason why the output power does not vary in the region Y.
Thus, because of the three regions X, Y, and Z, the transmitter shown in FIG. 11 has a narrow dynamic range (hereinafter this problem will be referred to as “the third problem”).
As wireless communication increases in speed and in capacity in the years to come, transmission circuits will face even severer requirements as to being able to amplify a modulated signal over a wide dynamic range spanning from a low output level to a high output level, with a high efficiency and free of distortion. As described above, the transmitter shown in FIG. 11 has a third problem of having a narrow dynamic range. In addition, the transmitter shown in FIG. 11 has a fourth problem in that the dynamic range becomes even narrower as a higher modulation speed is used, for the following reason. The amplitude modulation signal Efm(t) is split into two portions so as to be input to the power supply section of the inter-stage amplifier 935 and to the power supply section of the last-stage amplifier 936. However, as the modulation speed increases, there emerges a discrepancy between the timing with which the amplitude of the output signal from the inter-stage amplifier 935 varies, and the timing with which the last-stage amplifier 936 performs amplitude modulation, thus making the dynamic range even narrower.
In view of the first and second problems above, a first object of the present invention is to provide an amplifier for transmission signals incorporating a “large-sized” transistor, such that it is possible to attain a gain which is relatively high and is uniform over a wide over a wide frequency range.
In view of the third and fourth problems above, a second object of the present invention is to provide a small-sized and inexpensive amplifier or transmitter capable of performing a polar coordinate modulation operation, such that a modulated signal can be amplified with a high efficiency and free of distortion over a wide dynamic range, without requiring complicated control.